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Spoof Surface Plasmon-Based Leaky-Wave Antenna (LWA)

  • Amin Kianinejad
Chapter
Part of the Springer Theses book series (Springer Theses)

Abstract

In this chapter, the spoof surface plasmon modes are implemented to design a single-layered leaky-wave antenna (SL-LWA). With a simple and single layer configuration, the proposed design offers all the advantageous features of the conventional leaky-wave antennas such as frequency scanning beam, forward, broadside and backward radiations as well as broadband operation for broadside radiation. With very low non-radiative power at the end of the antenna, the proposed leaky-wave antenna does not require any loading termination. In the next section, the radiation from the proposed single-layered leaky-wave antenna is studied. In Section II, a simple design procedure for the SL-LWA is proposed. In Section III, the radiation performance of the antenna is experimentally evaluated.

Antenna radiation patterns Beam steering Leaky wave antennas (LWAs) Periodic structures Plasmons 

Radiation from travelling wave structures was first proposed by Hansen [1]. With their non-resonant nature, travelling wave antennas offer unique features such as high gain, wide operation bandwidth, and frequency scanning beam for applications in millimeter wave and microwave systems [2, 3, 4].

Usually, leaky-wave antennas are terminated by a broadband load to absorb the non-radiated power and prevent it from reflecting back to the antenna and disturbing desired radiation patterns. This termination reduces the antenna efficiency, especially at higher frequencies where non-radiated power increases. H. V. Nguyen et al. proposed a power recycling feedback system to solve this issue for maximizing the antenna efficiency [5]. However, this additional power recycling unit increases the total size as well as the total ohmic loss of the antenna.

Among LWAs, the planar versions have been of more interest due to their low-profile configurations [6, 7, 8, 9, 10]. Recently, the planar LWAs designed by composite right/left handed (CRLH) structures have been proposed to realize a wide bandwidth of the consistent broadside gain (4.2%) [11]. This design as well as many other CRLH material based leaky-wave antennas require an arrangement of metal patches and via holes with two metallic layers.

Recently, spoof surface plasmon modes have been of interest in antenna engineering [12, 13, 14, 15, 16, 17, 18, 19, 20]. In [14], an array of microstrip patch antennas are fed by an SP-based waveguide. In [18], additional gaps are implemented between the cells of an SP-based transmission line to convert the slow waves to fast radiating waves, and design an SP-based leaky wave antenna with fixed radiation patterns.

In this chapter, the spoof surface plasmon modes are implemented to design a single-layered leaky-wave antenna (SL-LWA). With a simple and single layer configuration, the proposed design offers all the advantageous features of the conventional leaky-wave antennas such as frequency scanning beam, forward, broadside and backward radiations as well as broadband operation for broadside radiation. With very low non-radiative power at the end of the antenna, the proposed leaky-wave antenna does not require any loading termination. In the next section, the radiation from the proposed single-layered leaky-wave antenna is studied. In Sect. 5.2, a simple design procedure for the SL-LWA is proposed. In Sect. 5.3, the radiation performance of the antenna is experimentally evaluated.

5.1 Radiation Mechanism

The operation principle of the periodic leaky-wave antennas is based on generation of a radiating space harmonic [2]. Here, the operating leaky-wave mode of the SSP-based structures is employed to design a new type of leaky-wave antennas.

5.1.1 Single-layered Leaky-Wave Antenna

The meander-line slow wave transmission line (SW-TL) is composed of periodic meander SSP cells connected to CPW lines via two converters and is depicted in Fig. 5.1a. The meander SSP cell is presented in Fig. 5.1a inset with the geometrical parameters as follows: the period length is p, groove depth and groove width are d and g, respectively and the total width is l. The design procedure of the meander SW-TL is detailed in Sect.  3.2.3. The simulation results for the scattering parameters from the designed meander-line SW-TL are presented in Fig. 5.1b. The substrate is Rogers 4003 with the thickness of 1.5 mm and the loss tangent of 0.0027 and the geometrical parameters of the cell are p = 12 mm, d = 2 mm, g = p/2 and l = 5 mm. The cut off frequency for this cell is 9 GHz, as seen in Fig. 5.1b. Below 9 GHz, the meander-line SW-TL operates in the transmission line mode with transmission above −0.5 dB and reflection below −15 dB. For the frequencies range of 9–10.4 GHz transmission is suppressed to below −30 dB with high reflection, while above 10.4 GHz, reflection decreases with negligible transmission.
Fig. 5.1

Meander SW-TL. a The schematic of the design. Inset: the meander SSP cell. b Scattering parameter results from the meander SW-TL.

Figure adopted and reproduced with permission from Ref. [19]

Figure 5.2a depicts the dispersion curves for the two modes in the meander-line SW-TL. These results are calculated by taking the Fourier transform of the waves on the SW-TL at various frequencies. The space harmonic corresponds to the Bloch mode with n = −1 and the wavenumber (β−1):
$$ \beta_{ - 1} = \beta_{0} - 2\pi /p, $$
(5.1)
where β0 is the wavenumber of the dominant mode. The red dashed lines in Fig. 5.2a indicate the fast wave limit with lower wavenumber than the free space k0. For the frequency range of 10.4–24.5 GHz, the first space harmonic lies in the radiation range, where the reflection and the transmission are very low, according to the results in Fig. 5.1b. Therefore, the energy leaks from the transmission line and the SW-TL acts as a single-layered LWA (SL-LWA).
Fig. 5.2

Radiation from the SW-TL. a Dispersion curves for the dominant mode and the first space harmonic. b Normalized co-component (φ) radiation patterns of the SL-LWA in the y-z plane at frequencies. c 3D radiation pattern at 17 GHz.

Figure adopted and reproduced with permission from Ref. [19]

The main beam radiation direction θmax is calculated from the radiating space harmonic dispersion curve as below:
$$ \theta_{ \hbox{max} } = \left\{ \begin{aligned} \text{Arcsin} \left( {\frac{\beta - 1}{{k_{0} }}} \right) \hfill \\ 180^\circ - \text{Arcsin} \left( {\frac{\beta - 1}{{k_{0} }}} \right) \hfill \\ \end{aligned} \right.\quad ({\text{degree}}) $$
(5.2)

The first angle corresponds to radiation in the upper space (z > 0) while the second angle corresponds to that in the lower space (z < 0).

Figure 5.2b depicts the normalized radiation patterns of the SL-LWA for the co-component (φ) in the y-z plane. At 10.5 GHz the main beam is titled toward θmax = −120°. As the frequency increases from 10.5 to 17 GHz, the radiation direction changes from backward to broadside radiation. This is in accordance with the results in Fig. 5.2a, where the momentum of the radiating space harmonics increases from a negative value at 10.5 GHz to zero at 17 GHz. Figure 5.2c plots the three-dimensional radiation pattern at 17 GHz. By increasing the frequency from 17 to 24.5 GHz, the momentum increases and the radiation beam moves from broadside to forward direction (θmax = 30°). These results are in accordance with Eq. (5.2).

5.1.2 SL-LWA Without 2nd Converter

The scattering parameter results in Fig. 5.1b shows that the \( \left| {{\text{S}}_{21} } \right| \) of the SL-LWA within the radiation range from f = 10.4 to 24.5 GHz is lower than −12 dB. In other words, majority of the energy over the entire bandwidth radiates and the transmitted energy to Port 2 is almost zero. This is an advantage of the proposed design in comparison with other types of LWAs, where the transmitted power increases by increasing the frequency and the radiation efficiency of the antenna decreases significantly at higher frequencies.

The SL-LWA is composed of two converters as seem in Fig. 5.1a. To study the effect of the 2nd converter on the radiation, this part is removed (see Fig. 5.3a), and for the two cases of with and without the 2nd converter, the reflection spectrum and radiation performance are compared in Fig. 5.4a, b, respectively. Removing the 2nd converter does not affect the reflected power, radiation angle, and maximum gain. In other words, the majority of the electromagnetic energy has been radiated before reaching the 2nd converter in Fig. 5.1b while this part of antenna does not have a significant role in the radiation performance. Besides that, without any loading termination at the end of the SL-LWA, the performance of the antenna remains unchanged.
Fig. 5.3

Designed SL-SW-TL. a SL-LWA without the second converter. b The magnetic field distribution near the SL-LWA.

Figure adopted and reproduced with permission from Ref. [19]

Fig. 5.4

Effects of the 2nd mode converter. a The reflection spectrum, b the maximum radiation direction and the gain.

Figure adopted and reproduced with permission from Ref. [19]

5.1.3 Effect of Dielectric and Metal

To study the effect of the dielectric substrate on the radiation performance, the antenna without the substrate is simulated and the scattering parameter results are presented in Fig. 5.5a. Removing the substrate changes the impedance bandwidth to 12.7–35 GHz and increases the bandwidth from 80 to 94%. The second converter of the SL-LWA without dielectric is removed, similar to the configuration in Fig. 5.3a. Figure 5.5b shows the maximum radiation angle and the maximum gain of the antenna in the y-z plane and indicates that the SL-LWA without the supporting dielectric still operates as a leaky-wave antenna.
Fig. 5.5

Dielectric substrate effect of SL-SW-TL. a The scattering parameter results. b The maximum radiation angle and the gain.

Figure adopted and reproduced with permission from Ref. [19]

Figure 5.6a shows the total efficiency of the SL-LWA with and without the substrate. For the SL-LWA with substrate, three cases are studied with different lt (see Fig. 5.3a). For the SL-LWA without substrate, the dielectric loss is zero and the antenna efficiency is about 95% for the entire frequency range. This figure indicates that at 23.6 GHz, the efficiency decreases to 90%. At this frequency, the main beam points at θmax = 0° and the radiation is broadside as seen in Fig. 5.5b. The slight reduction of the antenna efficiency is due to the slight increase of the reflection at the broadside frequency, according to Fig. 5.5a. Although, the antenna efficiency is above 90% over the entire bandwidth showing the excellent radiation performance of the SL-LWA without the substrate.
Fig. 5.6

Efficiency of SL-LWA. a Total efficiency with and without the dielectric substrate. b The Radiation efficiency for three cases: SL-LWA (1) with lossy metal and dielectric, (2) with lossless dielectric and (3) with PEC.

Figure adopted and reproduced with permission from Ref. [19]

The efficiency of the SL-LWA with a 1.5-mm thick Rogers 4003 dielectric substrate layer reduces to 82%. The antenna efficiency reduces due to the additional losses introduced by the substrate. Moreover, according to the results in Fig. 5.6a, changing lt does not affect the total antenna efficiency significantly. Although, this parameter affects the antenna gain and is studied in Sect. 5.2. According to the results in this figure, the antenna efficiency with the dielectric substrate is changing over the frequencies; however, it is above 80% for the entire bandwidth.

To further study the effect of the omhic losses caused by dielectric and metal on the antenna performance, the radiation efficiency of the SL-LWA for three cases are studied in Fig. 5.6b:
  1. (1)

    the SL-LWA with lossy metal and on a lossy substrate layer tagged as “Lossy”;

     
  2. (2)

    the SL-LWA with lossy metal and on a lossless substrate layer tagged as “Lossless substrate”;

     
  3. (3)

    the SL-LWA with PEC as metal and on a lossy substrate layer tagged as “PEC”.

     

These results demonstrate that the effect of the metal loss on the radiation efficiency is less than that by the dielectric loss. In addition, the both losses reduce the efficiency by about 1 dB over the entire bandwidth from 10.4 to 24 GHz.

Figure 5.7a shows the normalized attenuation constant of the fields of the SL-LWA for two cases of the lossy and lossless antennas. These results indicate that presence of the loss cause negligible effect on the field decay rate and radiation operation.
Fig. 5.7

Effect of the loss on SL-LWA. The normalized decay rate of the SL-LWA.

Figure adopted and reproduced with permission from Ref. [19]

5.1.4 Comparison with Other LW Structures Similar to SL-LWA

The SL-LWA is composed of the SSP cells as well as the flaring ground. The periodic cells generate the space harmonics and the radiating EM energy. The flaring ground converts the CPW modes to the SSP modes and does not have any significant effect on the radiation. Figure 5.3b shows the magnetic field distribution of the SL-LWA at 17 GHz and demonstrates that the majority of the electromagnetic energy is distributed near the meander strip, while the field distribution near the flaring ground is negligible.

To further study the effect of the grooves on the antenna operation, a structure similar to Fig. 5.3a without grooves is simulated, as shown in Fig. 5.8a. Figure 5.8b shows the reflection results for varying geometrical parameters. These results indicate that the flaring ground along with the central conducting strip without the grooves could not generate the wideband radiation for any of the parameter sets and therefore, this structure could not operate as a wideband antenna.
Fig. 5.8

A simple strip antenna without grooves. a The schematic. b The reflection results for varying geometrical parameters.

Figure adopted and reproduced with permission from Ref. [19]

The flaring ground of the SL-LWA may also resemble a tapered slot antenna (TSA) [21, 22]. While this part of the SL-LWA does not radiate, the flared slot in the TSAs radiates the electromagnetic energy and generates a fixed endfire beam. This pattern is different from the frequency scanning radiating beam of the SL-LWA. Their different radiation patterns indicate the major difference between the radiation mechanisms of these two types of antennas.

5.2 Design Procedure and Optimization

To change the operation frequency range of the SL-LWA, the dispersion curve of the radiating space harmonic is being tuned and shifted to the desired frequency ranges. By increasing and decreasing the periodicity, the dispersion line of this harmonic in Fig. 5.2b moves right and left, respectively. Figure 5.2b indicates that the SL-LWA operating at the dominant mode can be approximated by a medium with an effective permittivity of εeff = 2.2. Our simulation for SL-LWAs shows that above the cut off frequency and within the leaky-wave frequency range, the effective permittivity of the dominant mode only depends on the substrate permittivity and its thickness and does not change significantly by changing the geometrical parameters of the SSP cells. Therefore, according to Eq. (5.2), the space harmonic follows the following dispersion relation:
$$ \beta_{ - 1} = \sqrt {\varepsilon_{eff} } k_{0} - \frac{2\pi }{p}. $$
(5.3)
For the meander cell studied in Fig. 5.2b, p = 12 mm and the broadside radiation (β−1 ≃ 0) occurs at f = 17 GHz. According to Eq. (5.3), to set the broadside radiation at 14.5 and 20 GHz, the periodicities of p = 14 mm and p = 10 mm are required. Figure 5.9a, b studies the effects of the periodicity (p) on the operating bandwidth and radiation angle, respectively. As p increases, the operating frequency range shifts to lower edge while the bandwidth is kept as 80%. Consequently, the radiation angle curve shifts to lower frequencies while the scanning range remains constant. The broadside radiation frequency also corresponds to f = 14.5 and 20 GHz, as were calculated from Eq. (5.3).
Fig. 5.9

Effect of the period length on SL-LWA. a The reflection and b the maximum radiation angle with the varying period lengths.

Figure adopted and reproduced with permission from Ref. [19]

Figure 5.10a shows the maximum gain of the φ-component in the y-z plane for the three sets of parameters, where for d = 1.9 mm and w = 5.1 mm the most consistent as well as highest gain over the frequency are achieved.
Fig. 5.10

Realized gain of SL-LWA. The maximum realized gain in the y-z plane with varying a d and w. b Nm and lt. c The broadside realized gain.

Figure adopted and reproduced with permission from Ref. [19]

The other parameters to be tuned are the length of the converter (lm = Nm p) and lt, as tagged in Fig. 5.3a. Figure 5.9b plots the maximum gain of the φ-component in the y-z plane for varying lt and Nm. Among all the parameters studied in this figure, with the selection of Nm = 10 and lt = 80 mm chosen for prototyping the most consistent gain over the frequency range is achieved.

Figure 5.9c shows the broadside realized gain over the frequency range. This figure indicates that the operation band for less than 1-dB gain variation is 16.5–17.2 GHz or 4.2% of fractional bandwidth. Benefiting from a single layer and planar configuration, the proposed leaky-wave antenna offers its broadband operation with a consistent broadside gain comparable with previous designs.

5.3 Experimental Verification

The designed single-layered leaky-wave antenna was fabricated, as shown in Fig. 5.11a, and experimentally validated. The reflection from the antenna was measured using a vector network analyzer and compared with the simulation results in Fig. 5.11b and both the results agree well and demonstrate wide impedance bandwidth from 10.4 to 24.5 GHz. The measured \( \left| {{\text{S}}_{11} } \right| \) is slightly higher than the simulated values, probably caused by presence of the SMA connector that was not included in the simulation.
Fig. 5.11

Experimental evaluation of SL-LWA. a The fabricated prototype. b The measured reflection spectrum compared with the simulation results.

Figure adopted and reproduced with permission from Ref. [19]

The radiation patterns were measured in an anechoic chamber. As the operating frequency range of the power amplifier (PA) in the measurement set up was limited, the measurement was performed for two frequency ranges of 10.5–18 GHz and 18–24.5 GHz separately with two different PAs. For the higher frequency range, two low noise amplifiers were employed before the transmit antenna as well as after the antenna under the test to compensate for higher loss at higher frequencies and keep the measured results above the noise floor of the network analyzer.

The normalized radiation patterns in the y-z planes are compared with the simulation results in Fig. 5.12. More results are presented in Appendix B. The cross-polarized θ-component is at least 12 dB lower than the co-polarized φ-component at all the frequencies and all the angles. The experiment agrees the simulation and demonstrates the frequency scanning radiation of the SL-LWA. According to Fig. 5.2a, the momentum of the first space harmonic at 17 GHz is zero, corresponding to the broadside radiation, as seen in Fig. 5.12b. Below 17 GHz, β−1 is negative corresponding to the backward radiation as seen in Fig. 5.12a. As the frequency increases, β−1 increases and the main beam tilts to the forward direction, as seen in Fig. 5.12c, d.
Fig. 5.12

Normalized radiation pattern of SL-LWA. Comparison between the measured and simulated results for the co-component (φ) as well as cross-component (θ) in the y-z plane at a 10.5, b 17, c 20 and d 24 GHz.

Figure adopted and reproduced with permission from Ref. [19]

The gain in the maximum radiation direction at frequencies was measured and compared with the simulation in Fig. 5.13, where a good agreement between the results was observed. The gain variation across the entire bandwidth is limited to 2.5 dB. The constant gain is an important advantage of the proposed antenna over the conventional LWAs with dropping gain at the broadside and/or forward radiation. At 17 GHz, where the radiation is broadside, the gain is as high as the rest of the bandwidth and no gain reduction is observed.
Fig. 5.13

Maximum realized gain in y-z plane.

Figure adopted and reproduced with permission from Ref. [19]

5.4 Conclusion

A single-layered leaky-wave antenna based on meander SSP cells has been proposed and experimentally verified. The antenna has achieved a wide operating bandwidth of 80% with high and consistent total efficiency, gain, and wideband broadside radiation over 4.2% of fractional bandwidth. The reflection bandwidth has been up to 95%. With the unique simple design free from ground planes and via holes, the SL-LWAs open a vista for application of leaky-wave antennas in integrated microwave circuits.

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© Springer Nature Singapore Pte Ltd. 2018

Authors and Affiliations

  1. 1.Department of Electrical and Computer EngineeringNational University of SingaporeSingaporeSingapore

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