Protection of primary users in dynamically varying radio environment: practical solutions and challenges
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Abstract
One of the primary objectives of deploying cognitive radio (CR) within a dynamic spectrum access (DSA) network is to ensure that the legacy rights of incumbent licensed (primary) transmissions are protected with respect to interference mitigation when unlicensed (secondary) communications are simultaneously operating within the same spectral vicinity. In this article, we present noncontiguous orthogonal frequency division multiplexing (NCOFDM) as a promising and practical approach for achieving spectrally agile wireless data transmission that is suitable for secondary users (SUs) to access fragmented spectral opportunities more efficiently. Furthermore, a review of the current stateoftheart is conducted with respect to methods specifically designed to protect the transmissions of the primary users (PUs) from possible interference caused by nearby SU transceivers employing NCOFDM. These methods focus on the suppression of outofband (OOB) emissions resulting from the use of NCOFDM transmission. To achieve the required OOB suppression, we present two practical approaches that can be employed in NCOFDM, namely, the insertion of cancellation carriers and windowing. In addition to the theoretical development and proposed improvements of these approaches the computer simulation results of their performance are presented. Several realworld scenarios regarding the coexistence of both PU and SU signals are also studied using actual wireless experiments based on softwaredefined radio. These simulation and experimental results indicate that OOB suppression can be achieved under realworld conditions, making NCOFDM transmission a viable option for CR usage in DSA networks.
Keywords
Orthogonal Frequency Division Multiplex Cognitive Radio Cyclic Prefix Orthogonal Frequency Division Multiplex Symbol Orthogonal Frequency Division Multiplex Signal1 Introduction
The idea of cognitive radio (CR) encompasses opportunistic and dynamic access to spectrum resources that might be available at a certain location and time. These resources, called spectrum holes, especially in metropolitan areas, can potentially be fragmented with several noncontiguous spectral bands of different width. Moreover, the availability of these spectrum holes may dynamically change over time, as the licensed users (primary usersPUs) enter into and depart from a given location. There has been a substantial amount of research conducted with respect to finding suitable technologies capable of aggregating the available spectrum adaptively according to dynamics of spectrum holes availability, and to support the transmissions of the secondary users (SUs) in a spectrally efficient manner. In order to use the fragmented spectrum, an SU radio transceiver must be able to shape its emission to make best use of available resources while simultaneously respecting the incumbent spectral accessing rights of the PUs.
The key for achieving a spectrally agile waveform that enables the coexistence of both PU and SU transmissions within a specified spatial, temporal, and spectral vicinity is to exert strict control over the spectral extent of the transmitted signal. One spectrally agile waveform approach that has been receiving significant attention in recent years is noncontiguous orthogonal frequency division multiplexing (NCOFDM) [1, 2], which is based on the popular orthogonal frequency division multiplexing (OFDM) transmission technique. One of the primary advantages of using NCOFDM within the context of a dynamic spectrum access (DSA) network is that it provides the flexibility of deactivating, or nulling, specific subcarriers with zeros as input values such that there is no SU transmit power at frequency locations corresponding to the presence of PU emissions.
Despite its advantages, NCOFDM possesses several substantial technical issues that need to be resolved in order to make this form of wireless data transmission within a DSA environment a viable option. One of these issues is the shape of the NCOFDM spectrum outside of the intended transmission bandwidth, which is known to be relatively high when left untreated due to the Sinc pulse shapes of the individual databearing subcarriers. Consequently, if PU transmissions are located next to a collection of databearing subcarriers belonging to an NCOFDM signal, this may result in the former experiencing an unacceptable level of interference from the latter. Therefore, it is essential that the spectral shape of the NCOFDM waveform is treated such that the outofband (OOB) radiation is minimized.
In addition to the issue of OOB interference, OFDMbased waveforms are generally characterized by relatively high peaktoaveragepower ratio (PAPR), which makes the transmit signal vulnerable to nonlinear distortions, such as signal clipping in highpower amplifiers. If signal clipping does occur, the resulting transmission spectrum will broaden, thus yielding a potential interference situation with adjacent PU signals. Consequently, it is important to investigate suitable methods for reducing the PAPR of NCOFDM transceivers with the goal of mitigating OOB interference. There have been a number of articles dealing with this problem, suggesting either PAPR reduction methods (see an overview of these methods in [3] and the references therein), signal predistortion (e.g., [4]) or the linearization methods of a power amplifier. However, further development of these methods is required to make them sufficiently practical for the purposes of realizing transceiver implementations in actual realworld scenarios, such that the choice of an appropriate method should be able to handle the timevarying radio transmission environment, including dynamically changing types of the PU transmissions. Simultaneously, these methods should aim at achieving reasonable computational complexity, negligible performance degradation of the SU transmission, and low energy costs.
In this article, we present an investigation of spectrally agile waveforms based on NCOFDM and assess their suitability for achieving SU transmissions that are capable of respecting the rights of incumbent PU signals. In Section 2, we present an overview of NCOFDM transmission within the context of a cognitive radiobased DSA network. We then review in Section 3 existing methods for achieving flexible spectral waveforms using NCOFDM while simultaneously mitigating the effects of OOB interference. Section 4 provides a closer look at a promising technique for mitigating OOB interference that combines the insertion of socalled cancel lation carriers (CCs) with OFDM symbolbased windowing. Moreover, we present an enhanced optimization algorithm with reduced computational complexity and reduced energy costs. Finally, the proposed OOB interference reduction approach for NCOFDM is evaluated using actual wireless transceivers based on softwaredefined radio (SDR) technology within a controlled environment, and the results of these experiments are presented and discussed in Section 5.
2 Spectrally agile multicarrier waveform framework
A conventional wireless transmission system is usually allocated a specific frequency band for data communications. These wireless transmissions are usually licensed, which means they possess exclusive rights to the assigned frequency bands. Although much of the wireless spectrum up to 3 GHz has been assigned to licensed wireless applications, several measurements campaigns have shown that a substantial portion of the licensed frequency bands are underutilized across the temporal, spectral, and spatial domains [5]. To continue providing sufficient spectral bandwidth for satisfying both current and future wireless access needs, both spectrum policy makers and communication technologists have proposed an innovative approach with respect to the wireless spectrum usage via opportunistic spectrum access (OSA). Relative to traditional approaches for accessing spectrum, OSA allows for unlicensed wireless users to temporarily "borrow" unoccupied licensed frequency bands [6]. However, these unlicensed (i.e., secondary) devices must still guarantee interferencefree wireless access to incumbent licensed (i.e., primary) signals. In particular, it is essential that the OOB radiation generated by the SU wireless device is mitigated in order to prevent interference with PU wireless signals located in the frequency vicinity. Consequently, given this constraint on SU wireless transceivers, communication systems performing OSA require a level of spectral agility in order to operate in the presence of PU signals, especially when it comes to mitigating interference resulting from OOB radiation, as well as simultaneously transmitting across several unoccupied frequency bands that are fragmented across the wireless spectrum whose aggregate bandwidth satisfies the secondary transmission requirements.
Multicarrier modulation (MCM) possesses sufficient spectral agility in order to facilitate the transmission of data from unlicensed SU transmitters across several fragmented frequency bands simultaneously even in the presence of licensed PU signals, thus resulting in an increase in spectrum utilization [7]. In particular, subcarriers located in the frequency vicinity of unoccupied wireless spectrum can be used for transmitting data while those subcarriers that could potentially interfere with nearby PU signals can be deactivated or nulled. However, simply deactivating subcarriers for the purposes of OOB interference mitigation may not be sufficient for the neighboring PUs' interference tolerance levels. Moreover, in addition to achieving a required level of OOB interference within a given spectrum mask, an SU transmitter performing OSA must be capable of tailoring its spectral characteristics dynamically in order to avoid interference with the dynamically changing incumbent licensed PU transmissions. Finally, most MCM transmission approaches possess the possibility of exhibiting large envelope variations in the time domain that is often characterized by a high PAPR. This results from the combination of the subcarrier signals into a single composite multicarrier waveform in the time domain. When high PAPR occurs, the resulting transmission spectrum broadens and produces OOB interference regardless of whether the initial spectral waveform has been properly shaped at the transmitter for low OOB interference.
Overall, noncontiguous MCM techniques have been recognized as a suitable candidate for OSA due to their potential for achieving spectrally efficient communications by exploiting fragmented unoccupied spectrum while simultaneously achieving high data rates [8, 9]. In fact, this form of data transmission approach is wellsuited for future wireless communication systems, including CR systems [10]. As mentioned before, the NCOFDM scheme possesses the ability to efficiently use fragmented spectrum opportunities as well as perform spectrum shaping in order to suppress interference that may affect nearby primary wireless transmissions. To counteract the potential for significant OOB interference resulting from NCOFDM transmission, which can negatively affect neighboring wireless signals, several techniques have been proposed in the literature that are designed to significantly suppress these sidelobes in order to make coexistence between PUs and SUs feasible. On the other hand, the OOB reduction process can potentially increase the computational complexity and energy (power) utilization. Given the possible constraints of limited computational and energy resources available via a user equipment, a practical approach to this problem is needed that achieves a balance between the OOB interference mitigation efficiency and its associated costs.
3 OOB reduction techniques for spectrally agile multicarrier waveforms
The dilemma of how to mitigate the OOB interference in multicarrier systems has attracted substantial interest over the last decade. In this section, we present an overview of the major achievements in this field, and indicate two methods that are particularly attractive for the application in CR framework.
3.1 Stateofart techniques for OOB radiation reduction
The simplest method for achieving OOB interference reduction is to reserve a number of edge (guard) subcarriers (GS) to serve as a spectral buffer between PU and SU transmissions [7], i.e., deactivation of subcarriers. Although simple to implement, this method significantly decreases the spectral efficiency and does not provide sufficient OOB interference reduction in most scenarios.
Another approach to the OOB power reduction of an OFDM signal is to spectrally shape each individual subcarrier spectrum [7]. We will discuss this simple method called windowing (W) in the following subsection in greater detail. In the adaptive symbol transition (AST) method [11], similar to W, the timedomain samples in the transition region between consecutive symbols are chosen adaptively in order to minimize the OOB power. For the AST algorithm, the information about symbols mapped to each subcarrier is needed in order to assess the amount of OOB interference in the neighboring frequency bands. A meansquareerror (MSE) minimization method is used to determine the values of the timedomain samples in the transition region. The primary drawbacks of this method are high computational complexity and reduced throughput.
Another method, called constellation expansion (CE) [12], adjusts the modulated data symbols transmitted per subcarrier such that the OOB interference can be reduced while simultaneously not losing any data information or causing distortion. This is achieved by enlarging the modulation constellation and by allowing data symbols to be represented by any one of the two constellation points. As a result, the minimum distance between the constellation points is reduced, and the biterrorrate (BER) performance decreases.
Another method, called subcarriers weighting (SW) [13, 14], minimizes the signal OOB interference level by multiplying the data subcarriers by optimized real weighting coefficients. At the receiver, data symbols transmitted using the weighted subcarriers can be viewed as distorted, particularly for the high values of the weighting coefficients. Consequently, the authors suggest to impose a constraint on the weighting coefficients values. Simulation results exhibit significant OOB interference suppression. Some modifications to this method have been made in [15], where maximization of the channel capacity combined with OOB interference mitigation is addressed.
In the multiplechoice sequences (MCS) [16] method, for each sequence of data symbols to be transmitted in an OFDM symbol, a set of corresponding sequences representing it is calculated. The sequence yielding the lowest interference to adjacent bands is then chosen from this set and transmitted. To retrieve the initial data sequence at the receiver the identification number of the selected sequence has to be provided, what requires additional control channel for this sideinformation. A variant of the MCS method with reduced computational complexity is presented in [17]. In this method, the corresponding sequences are generated through the data symbols phases rotation of the multiple of π/ 2, and thus a limited number of possible sets of sequences must be examined to choose the optimum one. As the OFDM edge subcarriers possess the strongest influence on the OOB radiation, only those subcarriers are altered. Another variant of the MCS method involves its merging with other spectrum shaping algorithms, e.g., in [18] the authors combined the MCS method with both SW and CCs method.
Polynomial cancellation coding (PCC) has been proposed in [19] and revisited in [20]. This method not only reduces the OOB radiation but also lowers the OFDM signal sensitivity to phase and frequency errors. As neighboring subcarriers have firmly aligned spectra, the adjacent subcarriers are modulated with the same, appropriately scaled data symbol in order to reduce the sidelobes power. This is usually done for groups of two or three subcarriers. Although this method reduces the system throughput, this effect can be weakened as the cyclic prefix (CP) does not have to be added and coded redundancy can be used to increase SNR.
Another method for achieving OOB interference reduction, called spectral precoding (SP), has been described in [21, 22]. In this method, the correlation between the datasymbols transmitted on subcarriers is introduced by blockcoding. The codegenerating matrix is chosen so as to minimize the OOB radiation power. The SP method provides the lowest OOB interference levels relative to other methods simulated in [21]. On the other hand, it has been observed that the OOB interference suppression is not so high when the CP is applied.
Another method for reducing OOB interference, called extended active interference cancellation (EAIC) [23], is based on the insertion of special carriers that are designed to negatively combine with highpower sidelobes caused by the data subcarriers. The AIC subcarriers can be placed inside the adjacent transmission spectrum, usually at frequency locations that are nonorthogonal to the SU data subcarriers. The main drawback of this method results from this lack of orthogonality and thus, data symbols distortion. A variant of the EAIC method was presented in [24], where the sidelobe suppression approach was improved by using a long timedomain cancellation signal spanning over a number of consecutive OFDM symbols. This method results in an increase of BER due to increased interference relative to the method presented in [23]. In [25], this method is improved by introducing the constraint on the selfinterference power level.
An interesting approach to the mitigation of OOB interference, called partial response signaling (PRS) [26], makes the values on each subcarrier dependent on the subsequent OFDM symbols. This can be done by independent lowpass filters on each input of the inverse fast Fourier transform (IFFT) block. Although relatively substantial OOB interference suppression can be achieved even with very low order (23) filters, the reception of such a signal requires either a slicer or a Viterbi detector when treating PRS filtering after being influenced by the multipath propagation channel.
An observation that the OOB radiation is the result of the time domain noncontinuity between subsequent OFDM symbols was the basis for a spectrum shaping method presented in [27]. This method is called Ncontinuous OFDM (NC). The continuity of 0th to N thorder derivatives at the ends of the OFDM symbols is achieved by adding low power, complexvalued quantities to each active data subcarrier at the input of a IFFT block.
An entire class of methods that support the protection of PU signals from the effects of OOB interference is based on the use of power allocation schemes that not only maximize the throughput but also reduce the OOB interference power, e.g., refer to methods presented in [2, 28, 29]. However, as these approaches might be seen as part of radio resources management they will not be investigated here further.
Finally, the concept of modulated filterbanks (MFB) can be also successfully applied to suppress the sidelobes of the OFDM transmission [30]. MFB can be used for sidelobe suppression by applying them over the OFDM spectrum such that the series of bandpass filters allows only the required spectrum to pass through it while rejecting the unwanted OOB radiations in every subband.
3.2 Windowing
Referring to Figure 1, it is worth mentioning that to provide a relatively small throughput decrease, consecutive symbols overlapping with each other by β samples can yield an effective OFDM symbol duration of N + N_{CP} + β samples.
The primary advantages of this method is its low computational complexity, independence of the modulated data, and its suitability for NCOFDM. When employed by a CR communication system attempting to access the available spectrum in a dynamically varying radio environment, it is also important that the length and shape of the applied window can be also altered dynamically. This method is the most suitable for minimizing the interference in the PU transmission that is relatively distant in frequency from SU transmission band [7, 31]. The main drawback of this method is the decrease of throughput caused by the addition of the CS.
3.3 Cancellation carriers
where Π_{CC} is the maximum allowable power for CCs. Although the solution of (3) is widely known, and has been presented in [33, 34], the constraint (4) increases the computational complexity of the optimization problem significantly, requiring us to solve the Lagrange inequality for each OFDM symbol, which might become infeasible for wideband transmissions possessing a large number of subcarriers.
Another drawback of the CC method, apart from the computational complexity, is the linkperformance deterioration, i.e., an increase of the BER. This is due to the fact that an OFDM system usually operates under the total power constraint. If part of an OFDM symbol energy is sacrificed to the CCs, the remaining energy that can be used for data transmission is reduced, and this naturally results in an SNR loss and corresponding BER degradation.
Nevertheless, the CC algorithm has been extensively investigated, and a number of modifications and combinations of the CC algorithm with other methods has been presented in the literature, e.g., refer to proposed approaches in [35, 36, 37]. For example, active interference cancellation (AIC) [33] is a method similar to CCs solution, where in addition to the OFDM edgesubcarriers several other subcarriers inside the PU transmission band are also used to minimize the OOB radiation. However, as shown in the aforementioned paper, the AIC subcarriers inside the PU transmission band possess a negligible influence on the OOB interference. Moreover, they can significantly increase the computational complexity of the resulting implementation. The CCs method is very flexible in terms of defining the number of cancellation subcarriers and their power levels. Moreover, some of its shortcomings can be efficiently equalized if the W method with a parameterdefined window duration is also applied.
Summary of the main features of the OOB reduction methods for NCOFDM
Properties  GS  W  AST  CE  CCs*  SW  MCS  PCC  SP  EAIC  PRS  NC  MFB 

Increase of PAPR  No  No  No  No  Yes  No  No  No  No  Yes  No  No  Yes 
Computational complexity  Low  Low  High  High  High  High  High  Low  High  High  High  High  High 
Side information control channel  No  No  No  No  No  No  Yes  No  No  No  No  No  No 
Suitability for high order modulations  Yes  Yes  Yes  No  Yes  No  No  Yes  Yes  No  No  Yes  Yes 
Seemless reception  Yes  No  No  No  Yes  Yes  No  No  No  Yes  No  No  Yes 
Decrease of throughput  High  High  Low  No  Low  No  High  High  Low  Low  No  No  No 
Increase of BER  No  No  No  Yes  Yes  Yes  No  No  No  No  Yes  Yes  No 
4 Advances of the stateoftheart in the OOB power reduction: promising combination of windowing and CCs technique
4.1 Reducedcomplexity reducedpower combined CCs and windowing
The combination of CCs with windowing seems to be a promising spectrum shaping mechanism. While windowing method provides better OOB interference mitigation for spectrum components more distant from occupied OFDM band on the frequency axis, the CCs method has the same behavior for components closer to the OFDM nominal band as shown in [34]. Thus, the combination of both methods, which was also presented in [34], provides additional degrees of freedom as the number of CCs and window shapes can be altered to fulfill the transmission requirements. In this section, we present several additional enhancements to this combined approach, thus yielding a reduction in the computational complexity, reduction of the energyloss (energy inefficiency) due to the use of the CCs, and an improvement of the BER performance.
The system that we consider in this research consists of a conventional OFDM modulator, where the CCs unit, which performs the CCs algorithm, is employed prior to the Nsize IFFT block, and windowing is applied to the time domain signal after extending it with the CP. The resulting OFDMmodulated signal after the OOB interference reduction process is then fed to the digitaltoanalog converter and the IF/RF (Intermediate Frequency/Radio Frequency) frontend.
For a set of frequencysampling points l = {l_{1},...,l_{ δ }} defined in the optimization region, and for n ∈ c, the coefficients p_{n,l}are the elements of the matrix ${\mathbf{P}}_{\text{CC}}^{\left(\delta \times \gamma \right)}$, and can be precalculated. Similarly, for the data carriers, when n ∈ d, p_{n,l}defines the matrix ${\mathbf{P}}_{\text{DC}}^{\left(\delta \times \alpha \right)}$, and can be calculated offline.
where []^{+} denotes the pseudoinverse. Although such a solution is relatively fast with respect to computational complexity, since the multiplication of vector sd by a precalculated matrix is performed for each OFDM symbol, it suffers from several issues.
The reference system in this case is the one that employs the nulled guard sub carriers on the subcarriers used by the CCs method in the proposed system. Another significant drawback is a substantial increase of the PAPR that is caused by the high power values transmitted on the CCs correlated with the DCs. Apart from the PAPR value, usually the probability of peaks occurrence is also taken into account since it is conceivable that the timedomain peaks possessing moderate instantaneous power can cause nonlinear distortions and performance deterioration that can prove to be much worse than the high power (strong) peaks occurring relatively infrequently. On the basis of this observation, the PAPR is measured with a certain probability p PSPR. We will determine this metric later in this section when providing simulation results for a probability of p PAPR = 10^{3}.
where S(f) is the power spectral density (PSD) function of the considered NCOFDM secondaryuser signal, B_{SUCC} is the ba ndwidth of the considered NCOFDM secondaryuser transmission used by the data subcarriers (excluding cancellation subcarriers frequency bands), and f_{CC} is any one of the frequencies belonging to CCs bands. This definition for the SOR can be interpreted as the logarithm of the PSD peaks of CCs with respect to the mean power level in data carriers band. The occurrence of these peaks is measured with probability p_{SOR}. Note that in simulation results presented in the next subsection, p_{SOR} = 10^{1} will be considered. This probabilistic approach is required to take a varying characteristic of the PSD estimate into account.
where I^{(γ × γ)} is a γsize identity matrix, and W results from multiplication of the first two matrices in the above equation. Such an optimization has similar computational complexity to the optimization problem of (7), as only once for a given spectrum mask, and after the number of DCs and CCs are determined, the optimization (calculation of matrix W) is implemented. Then, for each OFDM symbol, matrixbyvector multiplication is carried out with precalculated matrix W elements. The performance and influence on various system parameters will be evaluated in the next section.
The optimization procedure described above significantly reduces the SNR loss typically found for a CCs method. This is obtained as a result of imposing a constraint on the value of the SOR, which consequently reduces the power assigned to CCs and increases power reserved for the DCs. Nevertheless, the reduced power available for data subcarriers still cause some deterioration of the reception quality. Therefore, we propose the following reception technique that makes use of the CCs inherent redundancy.
where ${\stackrel{\u0303}{\mathbf{s}}}_{\mathbf{d}+\mathbf{c}}$ is a received vertical vector at the output of FFT block containing distorted and noisy values of data and cancellation subcarriers. Although the calculation of matrix R can be quite complex, it needs to be performed only once for each channel instance and subcarrier pattern. Moreover, with a systematic code implementation, this method may be treated as optional, reserved only for high performance, high quality reception.
Note that the reference system for this definition, i.e., all subcarriers employed for data transmission, is prohibited from operating in the considered scenario, where the PU transmission protection is required and the SU sidelobes have to be reduced.
4.1.1 Simulation results
Below, we present the Monte Carlo simulation results using MATLAB and showing that our introduced modifications of the combined CC and W method improves the overall performance of the NCOFDM system in several ways. In our experiments, we assumed N = 256 subcarriers, where the subcarriers possessing the indices d = { 100,..., 62} ∪ {41,..., 11} ∪ {10,..., 40} ∪ {61,..., 101} are occupied by the QPSK data symbols, and there are three CCs placed on each side of data carriers blocks, i.e. c = { 103,  102, 101} ∪ { 10, 9, 8}∪{7, 8, 9}∪{41, 42, 43}∪{58, 59, 60}∪{102, 103, 104}. The subcarriers pattern of four data subcarrier blocks is separated with narrowband PUs, e.g., program making and special events (PMSE) devices such as professional wireless microphones with bandwidth of 200 kHz. Note that an explanation of the wideband and narrowband PU signals and scenarios under consideration with respect to the coexistence of the PU and SU transmissions are given in the next section, with the realworld experimental results. The duration of the CP equals N_{CP} = 16 samples, but the β = 16 samples of the Hanning window extension (equal to CS) are also used on each side of an OFDM symbol. The number of CCs and shaping window duration was chosen in such a way that the mean OOB interference power level is achieved at least 40 dB below the mean inband power level for reasonable value of μ, i.e., μ = 0.01. This OOB power attenuation is sufficient in order to respect several regulatory spectrum masks, e.g., IEEE802.11 g [39] or LTE userequipment [40] Spectrum Emission Mask (SEM).
It can be observed in Figure 4 that the OOB power attenuation decreases slowly with an increase of μ for small values of μ. Thus, when μ is low, there is no use in spending additional power on CCs since the spurious OOB emissions remain the same. On the other hand, lowpower CCs (for high μ values) do not provide improvement in OOB power over results obtained for windowing method without the application of CCs. However, the other metrics improve when μ increases. For example, the fluctuation of SOR ranges from 13.7 to 3.9 dB. It is worth mentioning that the rest of the system performance metrics are calculated with respect to the reference system, which does not use windowing and CCs for OOB power reduction. Instead, the CCs are replaced with zeros. The significant improvement is observed in PAPRincrease value that approaches zero, when μ becomes high. Both new optimization goal defined by (11), and proposed reception algorithm have influence on the values of an SNR loss with standard detection and with our proposed detection making use of the CCs redundancy. The stronger the limit is on the CCs power (the higher μ) the DCs power is not wasted as much on the CCs. Thus, the SNR loss changes from 4.8 dB for μ = 10^{6} to nearly 0 dB for μ = 10^{0}.
The results after employing our proposed detection method show that not only do the DC power levels reassigned to the CCs was recovered, but also an additional improvement was achieved thanks in part to the frequency diversity introduced by CCs treated as parity symbols of the block code. We observe that the coding gain for BER = 10^{4} (with respect to system without CCs) varies from 6.42 dB for μ = 4 × 10^{6} to 0.7 dB for μ = 1. For very low values of μ(μ < 4 × 10^{6}), the SNR loss caused by the introduction of CCs becomes higher than can be compensated for even by using high power CCs, which yields a coding gain decrease. The results presented in Figure 4 show that our reception algorithm making use of the CCs redundancy yields decent performance even in the assumed case of large fragmentation of available (not occupied by the PUs) frequency bands.
5 Realworld experimental results
5.1 Implementation setup
One application for the deployment of CR systems and DSA networks is the opportunistic spectral usage of unoccupied portions of the TV frequency bands by future mobile radio systems such as long term evolution (LTE) mobile radio communication [42]. A television whitespace (TVWS) is a region of wireless frequencies where several digital video broadcastingterrestrial (DVBT) channels are not used by a licensed transmission, and therefore can be temporarily borrowed by TVWS devices capable of operating in these bands as long as they respect the limits concerning the maximum allowable transmit power in this area, as well as the level of their OOB interference power.
Moreover, it is envisioned that wireless microphones will be operating in these TVWS regions and associated frequency bands, as well as other wireless devices used for PMSE [43]. Although several spectrum regulators anticipate reserving one TV channel for the exclusive access by PMSE equipment (e.g., U.K. Ofcom reserves channel 38), it is anticipated to be not sufficient for large events that commonly use over 100 wireless microphones. Hence, PMSE devices may be using other channels as well. In order to address this application scenario, we consider the PU signals in this scenario to consist of a DVBT transmission using an 8 MHz channel and a PMSE transmission possessing a 200 kHz bandwith. Moreover, the LTE transmissions are considered to be SU signals in these frequency bands. In particular, our tested system implements the LTElike transmission with N = 512 subcarriers, with the possibility of turning some of the subcarriers off, as well as using some of them as the CCs for OOB interference reduction. The subcarrier spacing is 15 kHz, and the useful spanned band is 7.68 MHz. The CP duration in samples N_{CP} = N/ 16 has been used, and the binary phase shift keying (BPSK) signalling has been applied at each subcarrier.
Note that for our SU system the PMSE band spans over 14 subcarriers, and therefore such a PMSE transmission is considered as narrowband PU signal. If we consider channelization of the available subcarriers in blocks of 16 subcarriers, one block of subcarriers has to be deactivated in order to protect such a narrowband PU signal when detected. Our second type of PU signal, the DVBT system uses at minimum 8 MHz channel, and thus more than the assumed SU bandwidth of 7.68 MHz. Therefore, it is considered to be a wideband PU signal, and if such a PU signal is detected, its channel must remain unoccupied by the SU system.
Employing the assumption that the LTE system is considered to be an SU signal, we investigate the OOB interference suppression taken from this system SEM. In particular, our goal is to achieve a 59 dB OOB interference power attenuation below the PSD level of data carriers for the first usecase of the highpower transmitter and call it usecase 1. Note, that the minimum required suppression defined in the LTE Base Station (BS) SEM employs this value, i.e., 59 dB. In our second usecase, called usecase 2, 26 dB will be the required OOB power attenuation, which is the typical value for the LTE user equipment transmitter that has to be obeyed in adjacent channels. Note that in order to protect various types of PU signals present in the spectrum, their required signaltointerference ratio must be considered together with the signal attenuation between the SU transmitter and the PU receiver. Moreover, the PU receive filters parameters and their sensitivity have to be taken into account.
It is envisioned that the TVWS geolocation databases will provide the information on the maximum inband and OOB power allowable at the specific location for the specific devices and services. Here, we assume that the PU signals and SU signals are located at a distance that allows the use of the standard SEMs for the reduction of the OOB interference. Nevertheless, our proposed shaping mechanism is designed so that it can fit flexibly to any existing SEM requirement and the OOB power suppression requirements can be changed dynamically when a new PU is detected in the adjacent channel or in the middle of transmission band of the SU.
In order to evaluate the flexibility of our spectrum shaping algorithms, we consider the following four scenarios of the PU and SU coexistence, namely:

Scenario 1: The SU system occupies continuous bandwidth, with only DC carrier turned off. The DVBT systems or densely located PMSE devices (the PUs) are detected to operate on both sides of the SU's band, which uses subcarriers of indices: {100, ..., 1}∪{1,..., 50}. The OOB power reduction mechanisms have to be used on both sides of the SU band.

Scenario 2: Outer wideband PUs (DVBT) are detected and one narrowband PU (PMSE device) in the middle of the SU transmission band (16 subcarriers turned off). The indices of the SU's used subcarriers are: {100,..., 8} ∪ {9,..., 50}.

Scenario 3: Outer wideband PUs (DVBT) are detected and two narrowband PU using and noncontiguous bands (PMSE devices) inside the SU's band. The indices of the SU's used subcarriers are: { 100,..., 8} ∪ {9,..., 50} ∪ {67,..., 100}.
The parameters of the proposed OOB power suppression method (CC and windowing) for the considered experimental scenarios and usecases (usecase 1 and usecase 2: 59 and 26 dB of the OOB power suppression, respectively)
Scenarios  Indices of used subcarriers  Usecase 1  Usecase 2  

β  γ _{ e }  μ  R _{ loss } [%]  β  γ _{ e }  μ  R _{ loss } [%]  
Scenario 1  {100,...,1} ∪ {1,...,50}  64  6  0.001  17.95  4  2  0.07  3.38 
Scenario 2  {100,...,8} ∪ {9,..., 50}  96  4  0.001  25  4  2  0.09  6.613 
Scenario 3  {100,...,8} ∪ {9,..., 50} ∪ {67,...,100}  96  4  0.001  27.07  4  2  0.09  7.78 
5.2 Experiment outcomes
In order to allow for the online reconfiguration, computationally efficient fast algorithm for optimization described in Section 4 has been applied, whose solution is given by formula (12). The most computationally complex operation, the matrix pseudoinverse, has been performed using CLAPACK [46] library. Other operations, i.e., matrixmatrix or matrixvector multiplication, have been performed using selfbuilt functions. However, their performance can be improved using low level specialized libraries such as BLAS. Our SU transmitter has been constructed to be fully reconfigurable, i.e., the indices of used data carriers, the numbers of CCs at each subcarriers block edge γ_{e}, or the window duration can be changed through the XML file.
Parameters of the experimental CR transmitter and the spectrum analyzer
Transmitter parameters  

The number of possible OFDM subcarriers N  512 
Cyclic prefix duration in samples  N/16 
Maximum bandwidth  5 MHz 
RF carrier frequency  2.46 GHz 
Spectrum analyzer parameters  
SPAN  5 MHz 
RBW  10 kHz 
VBW  30 kHz 
Number of averaging sweeps  100 
5.3 Complexity versus interference suppression performance
The complexity of the proposed combination of CCs and a windowing algorithm is evaluated in this subsection. At the transmitter side, the complexity is significantly reduced in comparison with the standard implementation of the CC method. It is achieved via precomputation of the W^{(γ × α}) matrix, which has the indirect constraint on the computational complexity embedded. Thus, the optimization procedure (which is particulary complex) does not have to be performed online for every OFDMsymbol. Instead it is implemented by multiplication of the data vector by the precomputed matrix, as described in Section 4. The most complex operation is the computation of the pseudoinverse of a δ +γ × γ complex matrix. To achieve required accuracy of this task it is done by means of singular value decomposition whose complexity is 26γ^{3} + 8δγ^{2} + 4δ^{2}γ [47]. As γ is a number of CCs, which is usually much lower than the number of sampling points 8 in the optimization region and lower than the number of DCs α, this operation is feasible for the implementation even for portable devices.
The most important in our implementation was to reduce the number of computations needed for each OFDM symbol. This aim was achieved as calculation of CCs requires only αγ complex multiplications and (α  1)γ additions. The second part of shaping is achieved by windowing that requires extending each OFDM symbol. On each side of this symbol β samples are multiplied by windowing slopes, and thus 2β complexbyreal multiplications are required. As subsequent OFDM symbols overlap by β samples, β pairs of complex numbers have to be added.
The proposed receiver discussed in Section 4 introduces some additional complexity over the standard OFDM reception. This proposed reception algorithm results in noticeable performance improvement compensating for the powerloss (power wastage) related to the use of CCs. Nevertheless it is optional, and plausible to be used by more powerful devices where the highest quality is required even at the cost of reasonably increased computational complexity. In such a case, the matrixbyvector multiplication using α(α + γ  1) additions and (α + γ)α multiplications is required for each OFDM symbol. The calculation of the reception (decoding) matrix is more complex as it requires computation of the pseudoinverse of the dimension (α + γ) × α complex matrix. Based on [47], one can evaluate the complexity of such an operation as 4(α + γ)^{2}α + 22α^{3} complex operations. In case of a slowly varying channel, the update of this matrix can be performed in long periodic time intervals. Efficient algorithms for updating the pseudoinverse on the basis of a previously computed pseudoinverse also exist, and can be used for the improved reception of the NCOFDM signal with CCs.
Computational complexity (the number of complex additions and multiplications) of the proposed spectrum shaping algorithm with reduced complexity (per an OFDM symbol)
Operations  The number of at the transmitter related to  The number of operations at the receiver operating in operations  

CC method implementation  Windowing  Overall spectrum shaping  Standard configuration  Proposed configuration  
Complex additions  (α 1)γ  β  (α  1)γ + β  0  α(α + γ  1) 
Complex multiplications  αγ  2β  αγ + 2β  α  α(α + γ) 
Total no. operations for:  
Scenario 1, Usecase 1  3300  192  3492  138  41262 
Scenario 2, Usecase 1  3792  288  4080  119  32011 
Scenario 3, Usecase 1  7320  288  7608  153  54009 
Scenario 1, Usecase 2  1164  12  1176  146  43654 
Scenario 2, Usecase 2  2024  12  2036  127  34163 
Scenario 3, Usecase 2  3756  12  3768  157  52909 
The receiver complexity in our system depends strongly on the applied configuration. Our proposed decoding method decreases the BER significantly, but is approximately 300 times more computationally complex than standard detection. In addition, matrixbased reception depends strongly on number of DCs, so the worst result is observed in scenario 3. Nevetheless our proposed decoding is optional, as explained previously, and thus, it can be used only by more powerful devices.
6 Conclusion
In this article, practical considerations with respect to the protection of primary user transmissions using CR systems performing DSA have been presented. In particular, we have identified NCOFDM technique as a viable transmission technology capable of supporting spectrally agile communications within the context of DSA. We have also presented several methods for handling the OOB interference generated by NCOFDM SU transmissions, observing that this type of interference can be reduced down to a level required to protect the transmission quality of the PU signals.
Moreover, we have identified the combination of the CCs and windowing methods to be particularly efficient in terms of the OOB interference power reduction, an in terms of its flexibility to adjust the SU spectrum to the required spectrum emission mask. Our proposed improvements and modifications have resulted in a reduction in the computational complexity of the energy consumption of the OOB interference reduction methods, regaining the power "wasted" by the CCs at the receiver by making use of the CCs correlation with the data carriers. The socalled spectrum overshooting problem has been also solved with our new proposed optimization technique. Simulation results show that when the optimization parameter μ is properly chosen, the performance metrics: PAPR, SOR, SNR_{loss} and OOB power attenuation reach their wellbalanced values.
Finally, our proposed reducedcomplexity, reducedpower OOB interference power reduction method was evaluated in several realworld scenarios involving PU and SU signals and in two basic usecases of the required spectrum emission masks. We have considered both narrowband (e.g., PMSE) and wideband (e.g., DVBT) PU signals in these realworld experiments, along with an LTElike SU transmission in TVWS with the possibility of aggregating the fragmented spectrum not used by the PU signals. The implementation has been conducted using an experimental SDR tested, showing exceptionally high degree of the frequency agility of our secondaryuser NCOFDM waveforms with the above presented novel practical design.
Endnote
^{a}This effect is caused by the USRP, not the spectrum analyzer, since the noise floor of the spectrum analyzer is much lower on each plot.
Notes
Acknowledgements
This study was supported in part by the European Commission, Seventh Framework Programme, under the project COGEU (contract no. ICT248560).
Supplementary material
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